The present invention relates to a switching power supply circuit to be provided as a power supply for various electronic apparatus.
Switching power supply circuits employing switching converters such as flyback converters and forward converters are widely known. These switching converters form a rectangular waveform in switching operation, and therefore there is a limit to suppression of switching noise. It is also known that because of their operating characteristics, there is a limit to improvement of power conversion efficiency.
Hence, there have been proposed various switching power supply circuits formed by various resonance type converters that make it possible to readily obtain high power conversion efficiency and to achieve low noise by forming a sinusoidal waveform in switching operation. The resonance type converters have another advantage of being able to be formed by a relatively small number of parts.
FIGS. 9 and 10 are circuit diagrams each showing an example of a prior art switching power supply circuit employing a resonance type converter.
This voltage resonance type converter is externally excited, and a MOS-FET, for example, is used as a switching device Q1.
A capacitor Cr is connected in parallel with a drain and a source of the switching device Q1. Capacitance of the capacitor Cr and leakage inductance obtained at a primary winding N1 of an isolating converter transformer PIT form a voltage resonant circuit. The parallel resonant circuit performs resonant operation according to switching operation of the switching device Q1.
A clamp diode (so-called body diode) DD is connected in parallel with the drain and source of the switching device Q1. The clamp diode DD forms a path of clamp current that flows during an off period of the switching device.
The drain of the switching device Q1 is connected to an oscillating circuit 11 in a switching driver 10. An output of the drain supplied to the oscillating circuit 11 is used to variably control an on period of switching operation of the switching device Q1 in switching frequency control.
The switching device Q1 is driven for switching operation by the switching driver 10 which is formed by integrating the oscillating circuit 11 and a driving circuit 12, and the switching frequency of the switching device Q1 is variably controlled for the purpose of constant-voltage control. Incidentally, the switching driver 10 in this case is provided as a single integrated circuit, for example.
The switching driver 10 is connected to a line of rectified and smoothed voltage Ei via a starting resistance Rs. The switching driver 10 starts operation by being supplied with power supply voltage via the starting resistance Rs at the start of power supply, for example.
The oscillating circuit 11 in the switching driver 10 performs oscillating operation to generate and output an oscillating signal. The driving circuit 12 converts the oscillating signal into a driving voltage, and then outputs the driving voltage to a gate of the switching device Q1. Thus, the switching device Q1 performs switching operation according to the oscillating signal generated by the oscillating circuit 11. Therefore, the switching frequency and duty ratio of an on/off period within one switching cycle of the switching device Q1 are determined depending on the oscillating signal generated by the oscillating circuit 11.
The oscillating circuit 11 changes frequency fs of the oscillating signal on the basis of the level of a secondary-side direct-current output voltage E0 inputted via a photocoupler 30. The oscillating circuit 11 changes the switching frequency fs and at the same time, controls the waveform of the oscillating signal in such a manner that a period TOFF during which the switching device Q1 is turned off is fixed and a period TON during which the switching device Q1 is turned on is changed. The period TON is variably controlled on the basis of the level of a switching resonance pulse voltage V1 across the parallel resonant capacitor Cr.
As a result of such operation of the oscillating circuit 11, the secondary-side direct-current output voltage E0 is stabilized.
The isolating converter transformer PIT transmits switching output of the switching device Q1 to the secondary side of the switching power supply circuit.
As shown in FIG. 11, the isolating converter transformer PIT has an E-E-shaped core formed by combining E-shaped cores CR1 and CR2 made for example of a ferrite material in such a manner that magnetic legs of the core CR1 are opposed to magnetic legs of the core CR2. A gap G is formed in a central magnetic leg of the E-E-shaped core in a manner as shown in the figure, and a primary winding N1 and a secondary winding N2 are wound around the central magnetic leg in a state in which the windings are divided from each other by using a dividing bobbin B. Thus, a state of loose coupling at a required coupling coefficient, for example k≈0.85 is obtained between the primary winding N1 and the secondary winding N2, and because of the looseness of the coupling, a saturated state is not readily obtained.
The gap G can be formed by making the central magnetic leg of each of the E-shaped cores CR1 and CR2 shorter than two outer legs of each of the E-shaped cores CR1 and CR2.
As shown in FIGS. 9 and 10, an ending point of the primary winding N1 of the isolating converter transformer PIT is connected to the drain of the switching device Q1, while a starting point of the primary winding N1 is connected to the rectified and smoothed voltage Ei. Hence, the primary winding N1 is supplied with the switching output of the switching device Q1, whereby an alternating voltage whose cycle corresponds to the switching frequency of the switching device Q1 occurs in the primary winding N1.
The alternating voltage induced by the primary winding N1 occurs in the secondary winding N2 on the secondary side of the isolating converter transformer PIT. In FIG. 9, a secondary-side parallel resonant capacitor C2 is connected in parallel with the secondary winding N2, and in FIG. 10, a secondary-side series resonant capacitor C2 is connected in series with the secondary winding N2. Therefore leakage inductance L2 of the secondary winding N2 and capacitance of the secondary-side parallel or series resonant capacitor C2 form a resonant circuit. The resonant circuit converts the alternating voltage induced in the secondary winding N2 into a resonance voltage, whereby voltage resonance operation is obtained on the secondary side.
The power supply circuit is provided with the parallel resonant circuit to convert switching operation into voltage resonance type operation on the primary side, and the parallel or series resonant circuit to provide voltage resonance operation on the secondary side. In the present specification, the switching converter provided with resonant circuits on the primary side and the secondary side as described above is referred to as a xe2x80x9ccomplex resonance type switching converter.xe2x80x9d
A rectifying and smoothing circuit comprising a bridge rectifier circuit DBR and a smoothing capacitor C0 is provided on the secondary side of the power supply circuit, whereby a secondary-side direct-current output voltage E0 is obtained. In the power supply circuit of FIG. 9, full-wave rectifying operation is provided by the bridge rectifier circuit DBR on the secondary side. In this case, the bridge rectifier circuit DBR is supplied with the resonance voltage by the secondary-side parallel resonant circuit, and then generates the secondary-side direct-current output voltage E0 whose level is substantially equal to that of the alternating voltage induced in the secondary winding N2. In the power supply circuit of FIG. 10, two rectifier diodes D01 and D02 are connected in a manner shown in the figure, and therefore the rectifier circuit on the secondary side forms a voltage doubler rectifier circuit. Thus, the rectifier circuit on the secondary side provides a secondary-side direct-current output voltage E0 that has a level twice that of the alternating voltage obtained in the secondary winding N2.
The secondary-side direct-current output voltage E0 is also inputted to the oscillating circuit 11 in the switching driver 10 on the primary side via the photocoupler 30 insulating the primary side from the secondary side.
As for secondary-side operation of the isolating converter transformer PIT, mutual inductance M between inductance L1 of the primary winding N1 and inductance L2 of the secondary winding N2 becomes +M or xe2x88x92M, depending on polarity (winding direction) of the primary winding N1 and the secondary winding N2, a connected relation between the rectifier diodes D01 and D02, and change in polarity of the alternating voltage induced in the secondary winding N2.
For example, an equivalent of a circuit shown in FIG. 12A has a mutual inductance of +M, while an equivalent of a circuit shown in FIG. 12B has a mutual inductance of xe2x88x92M.
This will be applied to the secondary-side operation of the isolating converter transformer PIT shown in FIG. 9 or FIG. 10. When the alternating voltage obtained at the secondary winding N2 has a positive polarity, an operation that causes rectified current to flow in the bridge rectifier circuit DBR can be considered a +M operation mode, or forward operation, whereas when the alternating voltage obtained at the secondary winding N2 has a negative polarity, an operation that causes rectified current to flow in the bridge rectifier diode DBR can be considered a xe2x88x92M operation mode, or flyback operation. Every time the alternating voltage obtained at the secondary winding N2 becomes positive or negative, the operation mode of the mutual inductance becomes +M or xe2x88x92M, respectively.
With such a configuration, power increased by effects of the primary-side parallel resonant circuit and the secondary-side parallel resonant circuit is supplied to a load side, and accordingly the power supplied to the load side is increased as much, thereby improving a rate of increase of maximum load power.
This is achieved because the primary winding N1 and the secondary winding N2 are wound in a state in which the windings are divided from each other to be loosely coupled to each other in the isolating converter transformer PIT, and thereby a saturated state is not readily obtained.
Stabilizing operation of the circuit shown in FIG. 9 or FIG. 10 is as follows.
As described above, the oscillating circuit 11 in the switching driver 10 on the primary side is supplied with the secondary-side direct-current output voltage E0 via the photocoupler 30. The oscillating circuit 11 changes the frequency of the oscillating signal for output according to change in the level of the supplied secondary-side direct-current output voltage E0. This means an operation of changing the switching frequency of the switching device Q1. Thus, resonance impedance of the primary-side voltage resonance type converter and the isolating converter transformer PIT is changed, and accordingly energy transmitted to the secondary side of the isolating converter transformer PIT is also changed. As a result, the secondary-side direct-current output voltage E0 is controlled so as to remain constant at a required level. This means that the power supply is stabilized.
When the oscillating circuit 11 of the power supply circuit shown in FIG. 9 or FIG. 10 changes the switching frequency, the period TOFF during which the switching device Q1 is turned off is fixed and the period TON during which the switching device Q1 is turned on is variably controlled. Specifically, by variably controlling the switching frequency as an operation for constant-voltage control, the power supply circuit controls the resonance impedance for switching output, and at the same time, controls the conduction angle of the switching device within a switching cycle. This complex control operation is realized by a single control circuit system.
FIGS. 13A, 13B, 13C, 13D, 13E, and 13F are waveform diagrams showing operation of the primary-side voltage resonance type converters in the power supply circuits shown in FIG. 9 and FIG. 10. FIGS. 13A, 13B, and 13C each show operation of the primary-side voltage resonance type converters at an alternating input voltage VAC=100 V and a maximum load power Pomax=200 W. FIGS. 13D, 13E, and 13F each show operation of the primary-side voltage resonance type converters at an alternating input voltage VAC=100 V and a minimum load power Pomin=0 W, or no load.
When the switching device Q1 performs switching operation, the primary-side parallel resonant circuit performs resonant operation during the period TOFF during which the switching device Q1 is turned off. Thus, as shown in FIGS. 13A and 13D, the parallel resonance voltage V1 across the parallel resonant capacitor Cr forms a sinusoidal resonance pulse waveform during the period TOFF.
The parallel resonant operation performed during the period TOFF causes a parallel resonance current Icr to flow through the parallel resonant capacitor Cr so as to form a substantially sinusoidal waveform and change from a positive direction to a negative direction during the period TOFF, as shown in FIGS. 13C and 13F.
A comparison of FIG. 13A with FIG. 13D indicates that the switching frequency fs is controlled so as to rise as load power Po is decreased, and the switching frequency fs is changed while fixing the period TOFF at a constant length and changing the period TON, during which the switching device Q1 is turned on.
The voltage resonance type converters formed as shown in FIGS. 9 and 10 change the level of the parallel resonance voltage V1 according to variation in load power. For example, the parallel resonance voltage V1 is 550 V at a maximum load power Pomax=200 W. whereas the parallel resonance voltage V1 becomes 300 V at a minimum load power Pomin=0 W. This means that the parallel resonance voltage V1 has a tendency to rise as the load power becomes heavier.
As shown in FIGS. 13B and 13E, a switching output current IQ1 flowing through the drain or the collector of the switching device Q1 is at a zero level during the period TOFF, and flows in a manner shown by the waveforms of FIGS. 13B and 13E during the period TON. The level of the switching output current IQ1 also has a tendency to rise as the load power Po becomes heavier. For example, according to FIGS. 13B and 13E, the switching output current IQ1 is 3.8 A at a maximum load power Pomax=200 W, whereas the switching output current IQ1 is 1 A at a minimum load power Pomin=0 W.
As characteristics of the power supply circuits shown in FIGS. 9 and 10, FIG. 14 shows characteristics of variations in the switching frequency fs, the period TOFF and the period TON within a switching cycle, and the parallel resonance voltage V1 with respect to the alternating input voltage VAC at a maximum load power Pomax=200 W.
As shown in the figure, the switching frequency fs is changed within a range of fs=110 KHz to 140 KHz for the alternating input voltage VAC=90 V to 140 V. This means an operation of stabilizing variation in the secondary-side direct-current output voltage E0 according to variation in direct-current input voltage. As for variation in the alternating input voltage VAC, the switching frequency is controlled so as to rise as the level of the alternating input voltage VAC is increased.
As for the period TOFF and the period TON within one switching cycle, the figure shows that the period TOFF is constant, as contrasted with the switching frequency fs, whereas the period TON is shortened as the switching frequency fs is increased.
The parallel resonance voltage V1 also changes according to variation in the commercial alternating-current power VAC; the level of the parallel resonance voltage V1 rises as the alternating input voltage VAC is increased.
The power supply circuits as shown in FIGS. 9 and 10 configured to stabilize the secondary-side direct-current output voltage by the complex control method change the peak level of the parallel resonance voltage V1 according to load conditions and variation in the alternating input voltage VAC, as is shown in FIGS. 13A to 13F and FIG. 14. When the level of the alternating input voltage VAC as the 100-V commercial alternating-current power AC for example rises to 140 V under conditions of a heavy load approximating the maximum load power, in particular, the parallel resonance voltage V1 rises to 700 V at the maximum, as shown in FIG. 14.
Therefore, for a commercial alternating-current power AC of 100 V, products having a withstand voltage of 800 V need to be selected for the parallel resonant capacitor Cr and the switching device Q1, which are supplied with the parallel resonance voltage V1, while for a commercial alternating-current power AC of 200 V, products having a withstand voltage of 1200 V need to be selected for the parallel resonant capacitor Cr and the switching device Q1. This results in increases in size and cost of the parallel resonant capacitor Cr and the switching device Q1.
In addition, switching characteristics of the switching device are degraded as its withstand voltage becomes higher. Thus, a product having a high withstand voltage selected for the switching device Q1 as described above increases power loss due to switching operation, and also reduces power conversion efficiency.
In view of the problems described above, it is an object of the present invention to miniaturize and improve power conversion efficiency in the switching power supply circuit configured as a complex resonance type switching converter.
According to the present invention, there is provided a switching power supply circuit comprised as follows.
The switching power supply circuit according to the present invention comprises a switching means including a main switching device for interrupting an inputted direct-current input voltage for output, a primary-side parallel resonant capacitor for forming a primary-side parallel resonant circuit that converts operation of the switching means into voltage resonance type operation, and an insulating converter transformer having a primary-side winding and a secondary-side winding for transmitting an output of the switching means obtained in the primary-side winding to the secondary-side winding, the primary-side winding and the secondary-side winding being wound so as to be loosely coupled to each other at a required coupling coefficient.
The switching power supply circuit on the secondary side comprises a secondary-side resonant circuit formed by connecting a secondary-side resonant capacitor to the secondary-side winding, and a direct-current output voltage generating means for rectifying an alternating voltage inputted from the secondary-side winding and thereby providing a direct-current output voltage.
The switching power supply circuit is characterized by further including an active clamp means for clamping a primary-side parallel resonance voltage generated across the primary-side parallel resonant capacitor during an off period of the main switching device, the active clamp means including an auxiliary switching device driven for switching operation by self-oscillation driving, and a switching driving means for effecting control for constant voltage by driving the main switching device for switching operation such that switching frequency of the main switching device is variably controlled according to level of the direct-current output voltage and an on/off period of the main switching device within one switching cycle is changed.
The configuration on the primary side described above allows the active clamp means to clamp the parallel resonance voltage generated during the off period of the main switching device to thereby suppress the resonance voltage. Therefore, products having lower withstand voltage may be used for components such as the switching devices and the primary-side parallel resonant capacitor provided in the power supply circuit.
Moreover, the active clamp means is driven by a self-oscillation driving circuit of simple configuration that includes a driving winding formed by winding a wire of the primary-side winding of the insulating converter transformer. Therefore, the active clamp means requires only a small number of parts, and thus greatly contributes to miniaturization of the power supply circuit in particular.